Method for reduction of aliasing introduced by spectral envelope adjustment in real-valued filterbanks

ABSTRACT

The present invention proposes a new method for improving the performance of a real-valued filterbank based spectral envelope adjuster. By adaptively locking the gain values for adjacent channels dependent on the sign of the channels, as defined in the application, reduced aliasing is achieved. Furthermore, the grouping of the channels during gain-calculation, gives an improved energy estimate of the real valued subband signals in the filterbank.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a divisional of U.S. Ser. No. 11/859,521 filed 21Sep. 2007, which claims priority to U.S. Ser. No. 10/652,397, whichclaims priority to Swedish application 0202770-4, filed 18 Sep. 2002,the entirety of each of which are herein incorporated by this referencethereto.

TECHNICAL FIELD

The present invention relates to systems comprising spectral envelopeadjustment of audio signals using a real-valued subband filterbank. Itreduces the aliasing introduced when using a real-valued subbandfilterbank for spectral envelope adjustment. It also enables an accurateenergy calculation for sinusoidal components in a real-valued subbandfilterbank.

BACKGROUND OF THE INVENTION

It has been shown in PCT/SE02/00626 “Aliasing reduction using complexexponential modulated filterbanks”, that a complex-exponential modulatedfilterbank is an excellent tool for spectral envelope adjustment audiosignals. In such a procedure the spectral envelope of the signal isrepresented by energy-values corresponding to certain filterbankchannels. By estimating the current energy in those channels, thecorresponding subband samples can be modified to have the desiredenergy, and hence the spectral envelope is adjusted. If restraints oncomputational complexity prevents the usage of a complex exponentialmodulated filterbank, and only allows for a cosine modulated(real-valued) implementation, severe aliasing is obtained when thefilterbank is used for spectral envelope adjustment. This isparticularly obvious for audio signals with a strong tonal structure,where the aliasing components will cause intermodulation with theoriginal spectral components. The present invention offers a solution tothis by putting restraints on the gain-values as a function of frequencyin a signal dependent manner.

SUMMARY OF THE INVENTION

It is the object of the present invention to provide an improvedtechnique for spectral envelope adjustment.

In accordance with a first aspect of the invention, this object isachieved by an apparatus for spectral envelope adjustment of a signal,comprising: means for providing a plurality of subband signals, asubband signal having associated therewith a channel number k indicatinga frequency range covered by the subband signal, the subband signaloriginating from a channel filter having the channel number k in ananalysis filterbank having a plurality of channel filters, wherein thechannel filter having the channel number k has a channel response whichis overlapped with a channel response of an adjacent channel filterhaving a channel number k−1 in an overlapping range; means for examiningthe subband signal having associated therewith the channel number k andfor examining an adjacent subband signal having associated therewith thechannel number k−1 to determine, whether the subband signal and theadjacent subband signal have aliasing generating signal components inthe overlapping range; means for calculating a first gain adjustmentvalue and a second gain adjustment value for the subband signal and theadjacent subband signal in response to a positive result of the meansfor examining, wherein the means for calculating is operative todetermine the first gain adjustment value and the second gain adjustmentvalue dependent on each other; and means for gain adjusting the subbandsignal and the adjacent subband signal using the first and the secondgain adjusting values or for outputting the first and the second gainadjustment values for transmission or storing.

In accordance with a second aspect of the invention, this object isachieved by a method of spectral envelope adjustment of a signal,comprising: providing a plurality of subband signals, a subband signalhaving associated therewith a channel number k indicating the frequencyrange covered by the subband signal, the subband signal originating froma channel filter having the channel number k in an analysis filterbankhaving a plurality of channel filters, wherein the channel filter havingthe channel number k has a channel response which is overlapped with achannel response of an adjacent channel filter having a channel numberk−1 in an overlapping range; examining the subband signal havingassociated therewith the channel number k and for examining an adjacentsubband signal having associated therewith the channel number k−1 todetermine, whether the subband signal and the adjacent subband signalhave aliasing generating signal components in the overlapping range;calculating a first gain adjustment value and a second gain adjustmentvalue for the subband signal and the adjacent subband signal in responseto a positive result of the means for examining, wherein the means forcalculating is operative to determine the first gain adjustment valueand the second gain adjustment value dependent on each other; and gainadjusting the subband signal and the adjacent subband signal using thefirst and the second gain adjusting values or outputting the first andthe second gain adjustment values for transmission or storing.

In accordance with a third aspect of the invention, this object isachieved by a computer program having a program code for performing theabove method, when the computer program runs on a computer.

In accordance with a fourth aspect of the invention, this object isachieved by a method for spectral envelope adjustment of a signal, usinga filterbank where the filterbank comprises a real valued analysis partand a real valued synthesis part or where said filterbank comprises acomplex analysis part and a real valued synthesis part, where a lower,in frequency, channel and an adjacent higher, in frequency, channel aremodified using the same gain value, if the lower channel has a positivesign and the higher channel has a negative sign, so that a relationbetween subband samples of the lower channel and subband samples of thehigher channel is maintained.

The present invention relates to the problem of intermodulationintroduced by aliasing in a real-valued filterbank used for spectralenvelope adjustment. The present invention analyses the input signal anduses the obtained information to restrain the envelope adjustmentcapabilities of the filterbank by grouping gain-values of adjacentchannel in an order determined by the spectral characteristic of thesignal at a given time. For a real-valued filterbank e.g. a pseudo-QMFwhere transition bands overlap with closest neighbour only, it can beshown that due to aliasing cancellation properties the aliasing is keptbelow the stop-band level of the prototype filter. If the prototypefilter is designed with a sufficient aliasing suppression the filterbankis of perfect reconstruction type from a perceptual point of view,although this is not the case in a strict mathematical sense. However,if the channel gain of adjacent channels are altered between analysisand synthesis, the aliasing cancellation properties are violated, andaliasing components will appear audible in the output signal. Byperforming a low-order linear prediction on the subband samples of thefilterbank channels, it is possible to assess, by observing theproperties of the LPC polynomial, where in a filterbank channel a strongtonal component is present. Hence it is possible to assess whichadjacent channels that must not have independent gain-values in order toavoid a strong aliasing component from the tonal component present inthe channel.

The present invention comprises the following features:

-   -   Analysing means of the subband channels to assess where in a        subband channel a strong tonal component is present;    -   Analysing by means of a low-order linear predictor in every        subband channel;    -   Gain grouping decision based on the location of the zeros of the        LPC polynomial;    -   Accurate energy calculation for a real-valued implementation.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention will now be described by way of illustrativeexamples, not limiting the scope or spirit of the invention, withreference to the accompanying drawings, in which:

FIG. 1 illustrates a frequency analysis of the frequency range coveredby channel 15 to 24 of an M channel subband filterbank, of an originalsignal containing multiple sinusoidal components. The frequencyresolution of the displayed analysis is intentionally higher than thefrequency resolution of the used filterbanks in order to display wherein a filterbank channel the sinusoidal is present;

FIG. 2 illustrates a gain vector containing the gain values to beapplied to the subband channels 15-24 of the original signal.

FIG. 3 illustrates the output from the above gain adjustment in areal-valued implementation without the present invention;

FIG. 4 illustrates the output from the above gain adjustment in acomplex-valued implementation;

FIG. 5 illustrates in which half of every channel a sinusoidal componentis present;

FIG. 6 illustrates the preferred channel grouping according to thepresent invention;

FIG. 7 illustrates the output from the above gain adjustment in areal-valued implementation with the present invention;

FIG. 8 illustrates a block diagram of the inventive apparatus;

FIG. 9 illustrates combinations of analysis and synthesis filterbanksfor which the invention can be advantageously used.

FIG. 10 illustrates a block diagram of the means for examining from FIG.8 in accordance with the preferred embodiment; and

FIG. 11 illustrates a block diagram of the means for gain adjusting fromFIG. 8 in accordance with the preferred embodiment of the presentinvention.

DESCRIPTION OF PREFERRED EMBODIMENTS

The below-described embodiments are merely illustrative for theprinciples of the present invention for improvement of a spectralenvelope adjuster based on a real-valued filterbank. It is understoodthat modifications and variations of the arrangements and the detailsdescribed herein will be apparent to others skilled in the art. It isthe intent, therefore, to be limited only by the scope of the impendingpatent claims and not by the specific details presented by way ofdescription and explanation of the embodiments herein.

In the following description a real-valued pseudo-QMF is used comprisinga real-valued analysis as well as a real valued synthesis. It should beunderstood however, that the aliasing problem addressed by the presentinvention also appears for systems with a complex analysis and areal-valued synthesis, as well as any other cosine-modulated filterbankapart from the pseudo-QMF used in this description. The presentinvention is applicable for such systems as well. In a pseudo-QMF everychannel essentially only overlaps its adjacent neighbour in frequency.The frequency-response of the channels is shown in the subsequentfigures by the dashed lines. This is only for illustrative purposes toindicate the overlapping of the channels, and should not be interpretedas the actual channel response given by the prototype filter. In FIG. 1the frequency analysis of an original signal is displayed. The figureonly displays the frequency range covered by 15·π/M to 25 ·π/M of the Mchannel filterbank. In the following description the designated channelnumbers are derived from their low cross-over frequency, hence channel16 covers the frequency range 16 ·π/M to 17·π/M excluded the overlapwith its neighbours. If no modification is done to the subband samplesbetween analysis and synthesis the aliasing will be limited by theproperties of the prototype filter. If the subband samples for adjacentchannels are modified according to a gain vector, as displayed in FIG.2, with independent gain values for every channel the aliasingcancellation properties are lost. Hence an aliasing component will showup in the output signal mirrored around the cross-over region of thefilterbank channels, as displayed in FIG. 3. This is not true for ancomplex implementation as outlined in PCT/SE02/00626 where the output,as displayed in FIG. 4, would not suffer from disturbing aliasingcomponents. In order to avoid the aliasing components that causes severeintermodulation distortion in the output, the present invention teachesthat two adjacent channels that share a sinusoidal component as e.g.channel 18 and 19 in FIG. 1, must be modified similarly, i.e. the gainfactor applied to the two channels must be identical. This is hereafterreferred to as a coupled gain for these channels. This of course impliesthat the frequency resolution of the envelope adjuster is sacrificed, inorder to reduce the aliasing. However, given a sufficient number ofchannels, the loss in frequency resolution is a small price to pay forthe absence of severe intermodulation distortion.

In order to assess which channels should have coupled gain-factors, thepresent invention teaches the usage of in-band linear prediction. If alow order linear prediction is used, e.g. a second order LPC, thisfrequency analysis tool is able to resolve one sinusoidal component inevery channel. By observing the sign of the first predictor polynomialcoefficient it is easy to determine if the sinusoidal component issituated in the upper or lower half of the frequency range of thesubband channel.

A second order prediction polynomial

A(z)=1−α₁ z ⁻¹−β₂ z ⁻²  (1)

is obtained by linear prediction using the autocorrelation method or thecovariance method for every channel in the QMF filterbank that will beaffected by the spectral envelope adjustment. The sign of the QMF-bankchannel is defined according to:

$\begin{matrix}{{{sign}(k)} = \left\{ {\begin{matrix}\left( {- 1} \right)^{k} & {{{if}\mspace{14mu} \alpha_{1}} < 0} \\\left( {- 1} \right)^{k + 1} & {{{if}\mspace{14mu} \alpha_{1}} \geq 0}\end{matrix},{0 < k < M},} \right.} & (2)\end{matrix}$

where k is the channel number, M is the number of channels, and wherethe frequency inversion of every other QMF channel is taken intoaccount. Hence, it is possible for every channel to assess where astrong tonal component is situated, and thus grouping the channelstogether that share a strong sinusoidal component. In FIG. 5 the sign ofeach channel is indicated and hence in which half of the subband channelthe sinusoidal is situated, where +1 indicates the upper half and −1indicates the lower half. The invention teaches that in order to avoidthe aliasing components the subband channel gain factors should begrouped for the channels where channel k has a negative sign and channelk−1 has a positive sign. Accordingly the channel signs as illustrated byFIG. 5 gives the required grouping according to FIG. 6, where channel 16and 17 are grouped, 18 and 19 are grouped, 21 and 22 are grouped, andchannel 23 and 24 are grouped. This means that the gain values g_(k)(m)for the grouped channels k and k−1 are calculated together, rather thanseparately, according to:

$\begin{matrix}{{{g_{k}(m)} = {{g_{k - 1}(m)} = \sqrt{\frac{{E_{k}^{ref}(m)} + {E_{k - 1}^{ref}(m)}}{{E_{k}(m)} + {E_{k - 1}(m)}}}}},} & (3)\end{matrix}$

where E_(k) ^(ref) (m) is the reference energy, and E_(k)(m) is theestimated energy, at the point m in time. This ensures that the groupedchannels get the same gain value. Such grouping of the gain factorspreserves the aliasing cancellation properties of the filterbank andgives the output according to FIG. 7. Here it is obvious that thealiasing components present in FIG. 3, are vanished. If there is nostrong sinusoidal component, the zeros will nevertheless be situated ineither half of the z-plane, indicated by the sign of the channel, andthe channels will be grouped accordingly. This means that there is noneed for detection based decision making whether there is a strong tonalcomponent present or not.

In a real-valued filterbank, the energy estimation is notstraightforward as in a complex representation. If the energy iscalculated by summing the squared subband samples of a single channel,there is a risk of tracking the time envelope of the signal rather thanthe actual energy. This is due to the fact that a sinusoidal componentcan have an arbitrary frequency from 0 to the filterbank channel width.If a sinusoidal component is present in a filterbank channel it can havea very low relative frequency, albeit being a high frequency sinusoidalin the original signal. Assessing the energy of this signal becomesdifficult in a real-valued system since, if the averaging time is badlychosen with respect to the frequency of the sinusoidal, a tremolo(amplitude-variation) can be introduced, when in fact the signal energyactually is constant. The present invention teaches however, that thefilterbank channels should be grouped two-by-two given the location ofthe sinusoidal components. This significantly reduces thetremolo-problem, as will be outlined below.

In a cosine-modulated filterbank the analysis filters h_(k)(n) arecosine-modulated versions of a symmetric low-pass prototype filter p₀(n)as

$\begin{matrix}{{h_{k}(n)} = {\sqrt{\frac{2}{M}}{p_{0}(n)}\mspace{11mu} \cos \left\{ {\frac{\pi}{2\; M}\left( {{2k} + 1} \right)\left( {n - \frac{N}{2} - \frac{M}{2}} \right)} \right\}}} & (4)\end{matrix}$

where M is the number of channels, k=0, 1, . . . , M−1, N is theprototype filter order and n=0, 1, . . . , N. The symmetry of theprototype filter is assumed here to be with respect to n=N/2. Thederivations below are similar in case of half sample symmetry.

Given a sinusoidal input signal x(n)=Acos(Ωn+θ) with frequency 0≦Ω≦π,the subband signal of channel k≧1 can be computed to be approximately

$\begin{matrix}{{{v_{k}(n)} \approx {\frac{A}{\sqrt{2M}}P\begin{Bmatrix}{\Omega - \frac{\pi}{2\; M}} \\\left( {{2k} + 1} \right)\end{Bmatrix}\cos \begin{Bmatrix}{{\Omega \; {Mn}} + \frac{\pi}{4}} \\{\left( {{2k} + 1} \right) - \frac{N\; \Omega}{2} + \theta}\end{Bmatrix}}},} & (5)\end{matrix}$

where P(ω) is the real valued discrete time Fourier transform of theshifted prototype filter p₀(n+N/2). The approximation is good whenP(Ω+π(k+½)/M) is small, and this holds in particular if P(ω) isnegligible for |ω|≦π/M, a hypothesis underlying the discussion whichfollows. For spectral envelope adjustment, the averaged energy within asubband k might be calculated as

$\begin{matrix}{{{E_{k}(m)} = {\sum\limits_{n = 0}^{L - 1}{{v_{k}\left( {{m\; L} + n} \right)}^{2}{w(n)}}}},} & (6)\end{matrix}$

where w(n) is a window of length L. Inserting equation (5) in equation(6) leads to

$\begin{matrix}{{{E_{k}(m)} = {\frac{A^{2}}{4M}P\begin{Bmatrix}{\Omega - \frac{\pi}{2\; M}} \\\left( {{2k} + 1} \right)\end{Bmatrix}^{2}\begin{Bmatrix}{{W(0)} + {{W\left( {2\Omega \; M} \right)}}} \\{\cos \begin{pmatrix}{{2\Omega \; {MLm}} + \frac{\pi}{2}} \\{\left( {{2k} + 1} \right) + {\Psi (\Omega)}}\end{pmatrix}}\end{Bmatrix}}},} & (7)\end{matrix}$

where Ψ(Ω) is a phase term which is independent of k and W(ω) is thediscrete time Fourier transform of the window. This energy can be highlyfluctuating if ω is close to an integer multiple of π/M, although theinput signal is a stationary sinusoid. Artifacts of tremolo type willappear in a system based on such single real analysis bank channelenergy estimates.

On the other hand, assuming that π(k−½)/M≦Ω≦π(k+½)/M and that P(ω) isnegligible for |ω|≦π/M, only the subband channels k and k−1 have nonzerooutputs, and these channels will be grouped together as proposed by thepresent invention. The energy estimate based on these two channels is

$\begin{matrix}{{{{E_{k}(m)} + {E_{k - 1}(m)}} = {\frac{A^{2}}{4M}{S_{k}(\Omega)}\begin{Bmatrix}{{W(0)} + {ɛ_{k}(\Omega)}} \\{\cos \begin{pmatrix}{{2\Omega \; {MLm}} + \frac{\pi}{2}} \\{\left( {{2k} + 1} \right) + {\Psi (\Omega)}}\end{pmatrix}}\end{Bmatrix}}},{where}} & (8) \\{{{S_{k}(\Omega)} = {{P\left\{ {\Omega - {\frac{\pi}{2\; M}\left( {{2k} + 1} \right)}} \right\}^{2}} + {P\left\{ {\Omega - {\frac{\pi}{2\; M}\left( {{2k} - 1} \right)}} \right\}^{2}}}}{and}} & (9) \\{{ɛ_{k}(\Omega)} = {{{W\left( {2\; \Omega \; M} \right)}}{\frac{{P\begin{Bmatrix}{\Omega - \frac{\pi}{2\; M}} \\\left( {{2k} + 1} \right)\end{Bmatrix}^{2}} - {P\begin{Bmatrix}{\Omega - \frac{\pi}{2\; M}} \\\left( {{2k} - 1} \right)\end{Bmatrix}^{2}}}{S_{k}(\Omega)}.}}} & (10)\end{matrix}$

For most useful designs of prototype filters, it holds that S(Ω) isapproximately constant in the frequency range given above. Furthermore,if the window w(n) has a low-pass filter character, then |ε(ω)| is muchsmaller than |W(0)|, so the fluctuation of the energy estimate ofequation (8) is significantly reduced compared to that of equation (7).

FIG. 8 illustrates an inventive apparatus for spectral envelopeadjustment of a signal. The inventive apparatus includes a means 80 forproviding a plurality of subband signals. It is to be noted that asubband signal has associated therewith a channel number k indicating afrequency range covered by the subband signal. The subband signaloriginates from a channel filter having the channel number k in ananalysis filterbank. The analysis filterbank has a plurality of channelfilters, wherein the channel filter having the channel number k has acertain channel response which is overlapped with a channel response ofan adjacent channel filter having a lower channel number k−1. Theoverlapping takes place in a certain overlapping range. As to theoverlapping ranges, reference is made to FIGS. 1, 3, 4, and 7 showingoverlapping impulse responses in dashed lines of adjacent channelfilters of an analysis filterbank.

The subband signals output by the means 80 from FIG. 8 are input into ameans 82 for examining the subband signals as to aliasing generatingsignal components. In particular, the means 82 is operative to examinethe subband signal having associated therewith the channel number k andto examine an adjacent subband signal having associated therewith thechannel number k−1. This is to determine whether the subband signal andthe adjacent subband signal have aliasing generating signal componentsin the overlapping range such as a sinusoidal component as illustratedfor example in FIG. 1. It is to be noted here that the sinusoidal signalcomponent for example in the subband signal having associated therewithchannel number 15 is not positioned in the overlapping range. The sameis true for the sinusoidal signal component in the subband signal havingassociated therewith the channel number 20. Regarding the othersinusoidal components shown in FIG. 1, it becomes clear that those arein overlapping ranges of corresponding adjacent subband signals.

The means 82 for examining is operative to identify two adjacent subbandsignals, which have an aliasing generating signal component in theoverlapping range. The means 82 is coupled to a means 84 for calculatinggain adjustment values for adjacent subband signals. In particular, themeans 84 is operative to calculate the first gain adjustment value and asecond gain adjustment value for the subband signal on the one hand andthe adjacent subband signal on the other hand. The calculation isperformed in response to a positive result of the means for examining.In particular, the means for calculating is operative to determine thefirst gain adjustment value and the second gain adjustment value notindependent on each other but dependent on each other.

The means 84 outputs a first gain adjustment value and a second gainadjustment value. It is to be noted at this point that, preferably, thefirst gain adjustment value and the second gain adjustment value areequal to each other in a preferred embodiment. In the case of modifyinggain adjustment values, which have been calculated for example in aspectral band replication encoder, the modified gain adjustment valuescorresponding to the original SBR gain adjustment values are bothsmaller than the higher value of the original values and higher than thelower value of the original values as will be outlined later on.

The means 84 for calculating gain adjustment values therefore calculatestwo gain adjustment values for the adjacent subband signals. These gainadjustment values and the subband signals themselves are supplied to ameans 86 for gain adjusting the adjacent subband signals using thecalculated gain adjustment values. Preferably, the gain adjustmentperformed by the means 86 is performed by a multiplication of subbandsamples by the gain adjustment values so that the gain adjustment valuesare gain adjustment factors. In other words, the gain adjustment of asubband signal having several subband samples is performed bymultiplying each subband sample from a subband by the gain adjustmentfactor, which has been calculated for the respective subband. Therefore,the fine structure of the subband signal is not touched by the gainadjustment. In other words, the relative amplitude values of the subbandsamples are maintained, while the absolute amplitude values of thesubband samples are changed by multiplying these samples by the gainadjustment value associated with the respective subband signal.

At the output of means 86, gain-adjusted subband signals are obtained.When these gain-adjusted subband signals are input into a synthesisfilterbank, which is preferably a real-valued synthesis filterbank, theoutput of the synthesis filterbank, i.e., the synthesized output signaldoes not show significant aliasing components as has been describedabove with respect to FIG. 7.

It is to be noted here that a complete cancellation of aliasingcomponents can be obtained, when the gain values of the adjacent subbandsignals are made equal to each other. Nevertheless, at least a reductionof aliasing components can be obtained when the gain adjustment valuesfor the adjacent subband signals are calculated dependent on each other.This means that an improvement of the aliasing situation is alreadyobtained, when the gain adjustment values are not totally equal to eachother but are closer to each other compared to the case, in which noinventive steps have been taken.

Normally, the present invention is used in connection with spectral bandreplication (SBR) or high frequency reconstruction (HFR), which isdescribed in detail in WO 98/57436 A2.

As it is known in the art, spectral envelope replication or highfrequency reconstruction includes certain steps at the encoder-side aswell as certain steps at the decoder-side.

In the encoder, an original signal having a full bandwidth is encoded bya source encoder. The source-encoder produces an output signal, i.e., anencoded version of the original signal, in which one or more frequencybands that were included in the original signal are not included anymore in the encoded version of the original signal. Normally, theencoded version of the original signal only includes a low band of theoriginal bandwidth. The high band of the original bandwidth of theoriginal signal is not included in the encoded version of the originalsignal. At the encoder-side, there is, in addition, a spectral envelopeanalyser for analysing the spectral envelope of the original signal inthe bands, which are missing in the encoded version of the originalsignal. This missing band(s) is, for example, the high band. Thespectral envelope analyser is operative to produce a coarse enveloperepresentation of the band, which is missing in the encoded version ofthe original signal. This coarse spectral envelope representation can begenerated in several ways. One way is to pass the respective frequencyportion of the original signal through an analysis filterbank so thatrespective subband signals for respective channels in the correspondingfrequency range are obtained and to calculate the energy of each subbandso that these energy values are the coarse spectral enveloperepresentation.

Another possibility is to conduct a Fourier analysis of the missing bandand to calculate the energy of the missing frequency band by calculatingan average energy of the spectral coefficients in a group such as acritical band, when audio signals are considered, using a grouping inaccordance with the well-known Bark scale.

In this case, the coarse spectral envelope representation consists ofcertain reference energy values, wherein one reference energy value isassociated with a certain frequency band. The SBR encoder nowmultiplexes this coarse spectral envelope representation with theencoded version of the original signal to form an output signal, whichis transmitted to a receiver or an SBR-ready decoder.

The SBR-ready decoder is, as it is known in the art, operative toregenerate the missing frequency band by using a certain or allfrequency bands obtained by decoding the encoded version of the originalsignal to obtain a decoded version of the original signal. Naturally,the decoded version of the original signal also does not include themissing band. This missing band is now reconstructed using the bandsincluded in the original signal by spectral band replication. Inparticular, one or several bands in the decoded version of the originalsignal are selected and copied up to bands, which have to bereconstructed. Then, the fine structure of the copied up subband signalsor frequency/spectral coefficients are adjusted using gain adjustmentvalues, which are calculated using the actual energy of the subbandsignal, which has been copied up on the one hand, and using thereference energy which is extracted from the coarse spectral enveloperepresentation, which has been transmitted from the encoder to thedecoder. Normally, the gain adjustment factor is calculated bydetermining the quotient between the reference energy and the actualenergy and by taking the square root of this value.

This is the situation, which has been described before with respect toFIG. 2. In particular, FIG. 2 shows such gain adjustment values whichhave, for example, been determined by a gain adjustment block in a highfrequency reconstruction or SBR-ready decoder.

The inventive device illustrated in FIG. 8 can be used for completelyreplacing a normal SBR-gain adjustment device or can be used forenhancing a prior art gain-adjustment device. In the first possibility,the gain-adjustment values are determined for adjacent subband signalsdependent on each other in case the adjacent subband signals have analiasing problem. This means that, in the overlapping filter responsesof the filters from which the adjacent subband signals originate, therewere aliasing-generating signal components such as a tonal signalcomponent as has been discussed in connection with FIG. 1. In this case,the gain adjustment values are calculated by means of the referenceenergies transmitted from the SBR-ready encoder and by means of anestimation for the energy of the copied-up subband signals, and inresponse to the means for examining the subband signals as to aliasinggenerating signal components.

In the other case, in which the inventive device is used for enhancingthe operability of an existing SBR-ready decoder, the means forcalculating gain adjustment values for adjacent subband signals can beimplemented such that it retrieves the gain adjustment values of twoadjacent subband signals, which have an aliasing problem. Since atypical SBR-ready encoder does not pay any attention to aliasingproblems, these gain adjustment values for these two adjacent subbandsignals are independent on each other. The inventive means forcalculating the gain adjustment values is operative to derive calculatedgain adjustment values for the adjacent subband signals based on the tworetrieved “original” gain adjustment values. This can be done in severalways. The first way is to make the second gain adjustment value equal tothe first gain adjustment value. The other possibility is to make thefirst gain adjustment value equal to the second gain adjustment value.The third possibility is to calculate the average of both original gainadjustment values and to use this average as the first calculated gainadjustment value and the second calculated envelope adjustment value.Another opportunity would be to select different or equal first andsecond calculated gain adjustment values, which are both lower than thehigher original gain adjustment value and which are both higher than thelower gain adjustment value of the two original gain adjustment values.When FIG. 2 and FIG. 6 are compared, it becomes clear that the first andthe second gain adjustment values for two adjacent subbands, which havebeen calculated dependent on each other, are both higher than theoriginal lower value and are both smaller than the original highervalue.

In accordance with another embodiment of the present invention, in whichthe SBR-ready encoder already performs the features of providing subbandsignals (block 80 of FIG. 8), examining the subband signals as toaliasing generating signal components (block 82 of FIG. 8) andcalculating gain adjustment values for adjacent subband signals (block84) are performed in a SBR-ready encoder, which does not do any gainadjusting operations. In this case, the means for calculating,illustrated by reference sign 84 in FIG. 8, is connected to a means foroutputting the first and the second calculated gain adjustment value fortransmittal to a decoder.

In this case, the decoder will receive an already “aliasing-reduced”coarse spectral envelope representation together with preferably anindication that the aliasing-reducing grouping of adjacent subbandsignals has already been conducted. Then, no modifications to a normalSBR-decoder are necessary, since the gain adjustment values are alreadyin good shape so that the synthesized signal will show no aliasingdistortion.

In the following, certain implementations of the means 84 for providingsubband signals are described. In case the present invention isimplemented in a novel encoder, the means for providing a plurality ofsubband signals is the analyser for analysing the missing frequencyband, i.e., the frequency band that is not included in the encodedversion of the original signal.

In case the present invention is implemented in a novel decoder, themeans for providing a plurality of subband signals can be an analysisfilterbank for analysing the decoded version of the original signalcombined with an SBR device for transposing the low band subband signalsto high band subband channels. In case, however, the encoded version ofthe original signal includes quantized and potentially entropy-encodedsubband signals themselves, the means for providing does not include ananalysis filterbank. In this case, the means for providing is operativeto extract entropy-decoded and re-quantized subband signals from thetransmitted signal input to the decoder. The means for providing isfurther operative to transpose such low band extracted subband signalsin accordance with any of the known transposition rules to the high bandas it is known in the art of spectral band replication or high frequencyreconstruction.

FIG. 9 shows the cooperation of the analysis filterbank (which can besituated in the encoder or the decoder) and a synthesis filterbank 90,which is situated in an SBR-decoder. The synthesis filterbank 90positioned in the decoder is operative to receive the gain-adjustedsubband signals to synthesize the high band signal, which is then, aftersynthesis, combined to the decoded version of the original signal toobtain a full-band decoded signal. Alternatively, the real valuedsynthesis filterbank can cover the whole original frequency band so thatthe low band channels of the synthesis filterbank 90 are supplied withthe subband signals representing the decoded version of the originalsignal, while the high band filter channels are supplied with the gainadjusted subband signals output by means 84 from FIG. 8.

As has been outlined earlier, the inventive calculation of gainadjustment values in dependence from each other allows to combine acomplex analysis filterbank and a real-valued synthesis filterbank or tocombine a real-valued analysis filterbank and a real-valued synthesisfilterbank in particular for low cost decoder applications.

FIG. 10 illustrates a preferred embodiment of the means 82 for examiningthe subband signals. As has been outlined before with respect to FIG. 5,the means 82 for examining from FIG. 8 includes a means 100 fordetermining a low order predictor polynomial coefficient for a subbandsignal and an adjacent subband signal so that coefficients of predictorpolynomials are obtained. Preferably, as has been outlined with respectto equation (1), the first predictor polynomial coefficient of a secondorder prediction polynomial as defined in the equation (1) iscalculated. The means 100 is coupled to means 102 for determining a signof a coefficient for the adjacent subband signals. In accordance withthe preferred embodiment of the present invention, the means 102 fordetermining is operative to calculate the equation (2) so that a subbandsignal and the adjacent subband signal are obtained. The sign for asubband signal obtained by means 102 depends, on the one hand, on thesign of the predictor polynomial coefficient and, on the other hand, ofthe channel number or subband number k. The means 102 in FIG. 10 iscoupled to a means 104 for analysing the signs to determine adjacentsubband signals having aliasing-problematic components.

In particular, in accordance with the preferred embodiment of thepresent invention, the means 104 is operative to determine subbandsignals as subband signals having aliasing-generating signal components,in case the subband signal having the lower channel number has apositive sign and the subband signal having the higher channel numberhas a negative sign. When FIG. 5 is considered, it becomes clear thatthis situation arises for subband signals 16 and 17 so that the subbandsignals 16 and 17 are determined to be adjacent subband signals havingcoupled gain adjustment values. The same is true for subband signals 18and 19 or subband signals 21 and 22 or subband signals 23 and 24.

It is to be noted here that, alternatively, also another predictionpolynomial, i.e., a prediction polynomial of third, forth or fifth ordercan be used, and that also another polynomial coefficient can be usedfor determining the sign such as the second, third or forth orderprediction polynomial coefficient. The procedure shown with respect toequations 1 and 2 is, however, preferred since it involves a lowcalculation overhead.

FIG. 11 shows a preferred implementation of the means for calculatinggain adjustment values for adjacent subband signals in accordance withthe preferred embodiment of the present invention. In particular, themeans 84 from FIG. 8 includes a means 110 for providing an indication ofa reference energy for adjacent subbands, a means 112 for calculatingestimated energies for the adjacent subbands and a means 114 fordetermining first and second gain adjustment values. Preferably, thefirst gain adjustment value g_(k) and the second gain adjustment valueg_(k−1) are equal. Preferably, means 114 is operative to performequation (3) as shown above. It is to be noted here that normally, theindication on the reference energy for adjacent subbands is obtainedfrom an encoded signal output by a normal SBR encoder. In particular,the reference energies constitute the coarse spectral envelopeinformation as generated by a normal SBR-ready encoder.

Depending on the circumstances, the inventive method of spectralenvelope adjustment can be implemented in hardware or in software. Theimplementation can take place on a digital storage medium such as a diskor a CD having electronically readable control signals, which cancooperate with a programmable computer system so that the inventivemethod is carried out. Generally, the present invention, therefore, is acomputer program product having a program code stored on amachine-readable carrier, for performing the inventive method, when thecomputer-program product runs on a computer. In other words, theinvention is, therefore, also a computer program having a program codefor performing the inventive method, when the computer program runs on acomputer.

1. Apparatus for assessing an energy of a signal having audio subbandsignals generated by filtering the signal using an analysis filterbank,the filterbank having subband filters, adjacent subband filters of thefilterbank having transition bands overlapping in an overlapping range,comprising: means for examining an audio subband signal originating froma subband filter and an adjacent audio subband signal originating froman adjacent subband filter, to determine whether the audio subbandsignal and the adjacent audio subband signal have aliasing generatingsignal components in the overlapping range to obtain grouped audiosubband signals in response to a result of the examination indicatingthat the audio subband signal and the adjacent audio subband signal havealiasing generating signal components in the overlapping range; andmeans for calculating an energy estimate for an energy in the groupedadjacent audio subband signals.
 2. Method of assessing an energy of asignal having audio subband signals generated by filtering the signalusing an analysis filterbank, the filterbank having subband filters,adjacent subband filters of the filterbank having transition bandsoverlapping in an overlapping range, comprising: examining an audiosubband signal originating from a subband filter and an adjacent audiosubband signal originating from an adjacent subband filter, to determinewhether the audio subband signal and the adjacent audio subband signalhave aliasing generating signal components in the overlapping range toobtain grouped audio subband signals in response to a result of theexamination indicating that the audio subband signal and the adjacentaudio subband signal have aliasing generating signal components in theoverlapping range; and calculating an energy estimate for an energy inthe grouped adjacent audio subband signals.
 3. Digital storage mediumhaving stored thereon a computer program having a program code forperforming the method in accordance with claim 2, when the computerprogram runs on a computer.